1. Field of the Invention
The present invention relates to multiple channel power amplifiers and methods for amplifying multiple channel (e.g., stereo) signals. In preferred embodiments, the invention is a system including circuitry for amplifying left and right channel signals, and a set of three transducers driven by the amplification circuitry (e.g., a loudspeaker for each of left, right, and subwoofer channels).
2. Description of the Related Art
FIG. 1 is a diagram of conventional circuitry for amplifying the left and right channel signals (11L and 11R) comprising a stereo audio signal. Typically, the left channel signal 11L and right channel signal 11R are highly correlated, in the sense that a substantial amount of the power of each is due to frequency components (typically those having low frequency) for which a frequency component of the left channel is at least substantially "in phase" with a corresponding frequency component of the right channel. Referring to FIG. 1, the typical connection of audio loudspeaker 18L to left channel stereo power amplifier 14L (as shown in FIG. 1), and of audio loudspeaker 18R to right channel stereo power amplifier 14R (as shown in FIG. 1) results in superposition of in-phase power supply current demands by the amplifier circuits for the left and right channels, when highly correlated left 11L and right 11R signals are amplified by their respective power amplifiers 14L, 14R to drive loudspeakers 18L, 18R.
To appreciate this, consider the case that highly correlated voltage signals 11R and 11L each have a frequency component (of frequency "f") and these frequency components are in-phase (so that their positive peaks are aligned when plotted along a common time axis, as are the positive peaks shown at the left side of FIG. 1). In this case, during amplification of the in-phase positive peaks of the highly correlated voltage signals 11L, 11R by amplifiers 14L and 14R, the corresponding power supply current demands (ISL and ISR) by the respective power amplifiers (14L and 14R) are in phase. This places a high demand upon the power supply which provides the supply voltage Vs for amplifiers 14R and 14L. Specifically, simultaneous positive peaks of voltage signals 11R and 11L cause amplifiers 14R and 14L to draw simultaneous positive pulses of current (ISL and ISR) from the power supply terminals (zero amplitude or negative amplitude portions of voltage signals 11R and 11L cause amplifiers 14R and 14L to draw a small, positive DC current from the power supply). Thus, in response to in-phase, sinusoidal input signals 11R and 11L, the combined current (ISL+ISR) drawn by both of amplifiers 14L and 14R is a sequence of superposed positive pulses of current ISL+ISR (as represented in FIG. 2). As illustrated, there are high, cumulative current demands by amplifiers 14R and 14L during the first half of each cycle (the period of each cycle is 1/f, where "f" is the frequency of each of sinusoidal input signals 11L and 11R). Thus, the power supply must be designed to operate with high peak to average current.
Unfortunately, most stereo audio sources have highly correlated left and right signals, particularly at the lower frequencies where power levels tend to be the highest. This is due to the generally non-directional nature of low frequency portions of most audio source material, and is thus generally unavoidable.
Above-referenced U.S. patent application Ser. No. 09/023,095 discloses several embodiments of an improved stereo signal amplifier, each operable with a power supply having reduced requirements (relative to those of power supplies of the type needed to operate conventional stereo amplifiers such as that of FIG. 1). FIG. 3 is a schematic diagram of one such embodiment. The system of FIG. 3 has two signal channels: a left channel including amplifier 114L and inverting amplifier 120 (which is a phase conversion circuit which inverts the phase of its input); and a right channel including amplifier 114R. The "right" channel signal 111R is coupled by input coupling capacitor 112R, and amplified by power amplifier 114R. The output of amplifier 114R is coupled via output coupling capacitor 116R to drive transducer 118R (e.g., an audio loudspeaker). The "left" channel voltage signal 111L is coupled by input coupling capacitor 112L, and inverted by inverting amplifier 120. The inverted left channel voltage signal (identified by reference numeral 121 in FIG. 3) is then amplified by power amplifier 114L. The amplified current output of amplifier 114L is coupled via output coupling capacitor 116L to drive transducer 118L (e.g., an audio loudspeaker).
As indicated in FIG. 3, each of transducers 118R and 118L is "polarized" in the sense that it has an inverting terminal (-) and a non-inverting terminal (+). Accordingly, depending upon which terminal (of transducer 118L or 118R) is connected to system ground GND and which terminal is driven by the output signal, the resulting transducer output signal (sound wave 119L or 119R) will have either the same phase or the opposite phase as the phase of the amplified current signal which drives it (i.e., the same phase or the opposite phase as the current signal output from amplifier 114L or 114R). Hence, in the left channel (in which transducer 118L has an inverted connection relative to the connection of right channel transducer 118R), the output 119L of transducer 118L is opposite in phase to the signal asserted at the output of power amplifier 114L. However, the phase shift introduced by inverting amplifier 120 cancels that introduced by transducer 118L, so that output audio wave 119L is in phase with original input signal 111L. Hence, the overall "left channel" phase shift between input voltage signal 111L and audio output signal 119L is approximately zero, just as the overall "right channel" phase shift between input signal 111R and audio output signal 119R is also approximately zero. Accordingly, the FIG. 3 circuit maintains an overall stereo effect while avoiding simultaneous positive current demands from both channels upon the power supply which provides supply voltage Vs across each of amplifiers 114R and 114L.
FIG. 4 shows the peak current demands on the power supply for amplifiers 114R and 114L (in response to in-phase frequency components of highly correlated signals 11R and 11L which are identical to the frequency components of signals 11R and 11L described above with reference to FIG. 2). Due to the phase inversion of left input signal 111L, there is no peak positive current ISL drawn (by amplifier 114L) during the first half of each cycle (amplifier 114L draws only a small, positive DC current I.sub.DC from the power supply, and amplifier 114R draws a peak positive current ISR from the power supply, during the first half of each cycle), and there is no peak positive current ISR drawn (by amplifier 114R) during the second half of each cycle (amplifier 114R draws only a small, positive DC current I.sub.DC, and amplifier 114L draws a peak positive current ISL, from the power supply during the second half of each cycle). Since the peak current demands by the left channel amplifier 114L and the right channel amplifier 114R occur during different (i.e., opposing) half-cycles of the input signals, the power supply for the FIG. 3 system can have a simpler design than the power supply for the FIG. 1 system, in that the power supply for the FIG. 3 system can be designed to operate with lower peak to average current demand than that expected during operation of the FIG. 1 system (and in that the size requirements for the energy storage capacitors of the power supply for FIG. 3 are reduced since the peak current demand is reduced). It will be understood that since the phase shift provided by inverting amplifier 120 is done prior to power amplification of left signal 111L, the power supply requirements by such amplifier 120 add a negligible peak current demand which coincides with the peak current demand of the right channel power amplifier 114R.
FIG. 5 is a schematic diagram of another embodiment of an improved stereo signal amplifier (operable with a power supply having reduced requirements) disclosed in above-referenced U.S. patent application Ser. No. 09/023,095. The FIG. 5 system is identical to that of FIG. 3, except in that a further advantage is realized by replacing output coupling capacitors 116R and 116L of FIG. 3 with a single output decoupling capacitor 216. Decoupling capacitor 216 is connected between system ground GND (or another system reference node) and a common transducer node 218. The individual transducers 118L, 118R are connected between node 218 and the outputs of power amplifiers 114L, 114R via their appropriate inverting (5) and non-inverting (+) terminals as shown. When connected in this way, the highly correlated low frequency amplified output current components (identified as "IOL" in FIG. 5) flow between the transducers 118L, 118R, while the less correlated high frequency output current components ("IOH") are effectively conducted to system ground GND by decoupling capacitor 216 (which can be much smaller than the output coupling capacitors 116L, 116R due to the frequencies of the current components being coupled).
It is contemplated that the circuitry of FIG. 3 or FIG. 5 (possibly including the power supply, but excluding the transducers and input and output capacitors) will be implemented as part of a single integrated circuit (i.e., an integrated circuit in which power amplifiers 114L and 114R, inverting amplifier 120, and possibly also the power supply, are integrated into a single substrate).